This invention relates to a switching power supply circuit to be used as a power supply for various electronic apparatus.
Various power supply circuits having a resonance type converter on the primary side have been proposed by the assignee of the present application. One such power supply circuit is disclosed, for example, in Japanese Patent Laid-Open No. 2003-235259 (hereinafter referred to as Patent Document 1).
FIG. 44 shows the switching power supply circuit disclosed in Patent Document 1 which includes a resonance type converter. Referring to FIG. 44, the power supply circuit shown includes a switching converter. The switching converter is configured such that a partial voltage resonance circuit for performing a voltage resonance operation only upon turn-off during switching is combined with a separately excited current resonance type converter of a half bridge coupling scheme.
The switching power supply circuit shown in FIG. 44 is provided, for example, as a power supply of a printer apparatus. In the printer apparatus, for example, the load power exhibits a variation over a comparatively wide range from approximately 100 W or more to no load.
In the power supply circuit shown in FIG. 44, a common mode noise filter formed from two filter capacitors CL and a common mode choke coil CMC is connected to a commercial AC power supply AC.
Further, as a rectification smoothing circuit for producing a DC input voltage from the commercial AC power supply AC, a full wave rectification circuit formed from a bridge rectification circuit Di and a smoothing capacitor Ci is provided at the stage following the common mode noise filter.
The rectification output of the bridge rectification circuit Di is charged into the smoothing capacitor Ci, and as a result, a rectification smoothed voltage Ei (DC input voltage) having a level equal to that of an AC input voltage VAC is obtained across the smoothing capacitor Ci.
The current resonance type converter for receiving and switching the DC input voltage includes a switching circuit system wherein two switching devices Q1 and Q2, each formed from a MOS-FET, are connected to each other in a half bridge coupling scheme as seen in FIG. 44. Damper diodes DD1 and DD2 each formed from a body diode are individually connected in parallel to each other and in such directions as seen in FIG. 44 between the drain-source of the switching devices Q1 and Q2, respectively.
Further, a partial resonance capacitor Cp is connected in parallel between the drain-source of the switching device Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed from the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of a primary winding N1. The partial voltage resonance circuit performs a partial voltage resonance operation wherein it voltage resonates only when the switching devices Q1 and Q2 are turned off.
In the power supply circuit, in order to switching drive the switching devices Q1 and Q2, an oscillation and driving circuit 2 formed from, for example, a general-purpose IC is provided. The oscillation and driving circuit 2 includes an oscillation circuit and a driving circuit, and applies a driving signal (gate voltage) having a required frequency to the gates of the switching devices Q1 and Q2. Consequently, the switching devices Q1 and Q2 perform a switching operation such that they are turned on and off alternately at a required switching frequency.
An insulating converter transformer PIT (Power Isolation Transformer) transmits the switching outputs of the switching devices Q1 and Q2 to the secondary side.
In this instance, the primary winding N1 of the insulating converter transformer PIT is connected at one end thereof to a node (switching output point) between the source of the switching device Q1 and the drain of the switching device Q2 through a series connection with a primary side series resonance capacitor C1 so that the switching output can be obtained from the node. The primary winding N1 is connected at the other end thereof to the primary side ground as seen in FIG. 44.
In this instance, the series resonance capacitor C1 and the primary winding N1 are connected in series to each other. Thus, a primary side series resonance circuit for making the operation of the switching converter that of a current resonance type is formed from the capacitance of the series resonance capacitor C1 and the leakage inductance L1 of the primary winding N1 (series resonance winding) of the insulating converter transformer PIT.
From the description above, resonance operation of the current resonance type by the primary side series resonance circuit (L1–C1) and partial voltage resonance operation by the partial voltage resonance circuit (Cp//L1) described hereinabove can be obtained by the primary side switching converter shown in FIG. 44.
In particular, the power supply circuit shown in FIG. 44 adopts a form in which the resonance circuit for making the operation of the primary side switching converter that of the resonance type is combined with another resonance circuit. A switching converter of the type just described is hereinafter referred to as a composite resonance type converter.
While a description with reference to the drawings is omitted here, the insulating converter transformer PIT described above is structured such that it includes an EE type core formed by combining E type cores made of, for example, a ferrite material with each other. Further, the primary winding N1 and a secondary winding N2 are wound on an inner magnetic leg of the EE type core at winding portions provided divisionally for the primary and secondary sides.
Further, a gap of 1.5 mm or less is formed in the inner magnetic leg of the EE type core of the insulating converter transformer PIT. Consequently, a coupling coefficient of 0.75 or more is obtained between the primary winding N1 and the secondary winding N2.
For the secondary winding N2 of the insulating converter transformer PIT, a full wave rectification circuit is provided. The full wave rectification circuit is formed from a bridge rectification circuit formed from rectification diodes Do1 to Do4, and a smoothing capacitor Co.
Consequently, a secondary side DC output voltage Eo, which is a DC voltage having a level equal to that of the alternating voltage induced in the secondary winding N2, can be obtained as a voltage across the smoothing capacitor Co. The secondary side DC output voltage Eo is supplied as a main DC power supply to a main load (not shown), and is also branched and input as a detection voltage for constant voltage control to the control circuit 1.
The control circuit 1 outputs a control signal to the oscillation and driving circuit 2 in the form of a current or a voltage whose level is adjusted corresponding to that of the secondary side DC output voltage Eo.
The frequency of an oscillation signal produced by the oscillation circuit in the oscillation and driving circuit 2 is adjusted based on the control signal input from the control circuit 1 to adjust the frequency of the switching driving signal to be applied to the gates of the switching devices Q1 and Q2. Consequently, the switching frequency is adjusted. Since the switching frequency of the switching devices Q1 and Q2 is adjustably controlled in response to the level of the secondary side DC output voltage Eo in this manner, the resonance impedance of the primary side series resonance circuit is varied and also the energy to be transferred from the primary winding N1, which forms the primary side series resonance circuit, to the secondary side is varied. Further, at this time, the level of the secondary side DC output voltage Eo is also adjustably controlled. Consequently, constant voltage control for the secondary side DC output voltage Eo can be implemented.
It is to be noted that such constant voltage control for adjustably controlling the switching frequency to achieve stabilization as described above is hereinafter referred to as the “switching frequency controlling method”.
FIG. 46 is a waveform diagram illustrating the operation of part of the power supply circuit shown in FIG. 44. Referring to FIG. 46, waveforms on the left side indicate the operation when the load power Po is Po=150 W, and waveforms on the right side indicate the operation of the same portions when the load power Po is Po=25 W. As an input voltage condition, the AC input voltage VAC is fixed at VAC=100 V.
Referring to FIG. 46, the voltage V1 of a rectangular wave is a voltage across the switching device Q2 and indicates on and off timings of the switching device Q2. The period that the voltage V1 is at the 0 level is an on period in which the switching device Q2 conducts. Within the on period, switching current IQ2 having the waveform illustrated in FIG. 46 is supplied to the switching circuit system formed from the switching device Q2 and the clamp diode DD2. Further, the period that the voltage V1 is clamped at the level of a rectification smoothed voltage Ei is a period in which the switching device Q2 is off, and the switching current IQ2 has the zero level as seen in FIG. 46.
Further, though not shown, the voltage obtained across the other switching device Q1 and the switching current flowing to the switching circuit (Q1, DD1) have waveforms shifted by 180° in phase from the waveforms of the voltage V1 and the switching current IQ2, respectively. In short, as described above, the switching devices Q1 and Q2 perform the switching operation such that they are turned on and off alternately.
Further, the switching currents flowing to the switching circuits (Q1, DD1 and Q2, DD2) are composed to obtain a current having the waveform shown in FIG. 46, and the resulting current is used as the primary side series resonance current Io to flow to the primary side series resonance circuit (C1–N1(L1)).
Further, it can be recognized, for example, from a comparison between the waveform of the voltage V1 shown in FIG. 46 when the load power Po=150 W and the waveform of the voltage V1 shown in FIG. 46 when the load power Po=25 W that the switching frequency is controlled. The switching frequency on the primary side, when the load to the secondary side DC output voltage Eo is heavy (Po=150 W), is lower than that when the load to the secondary side DC output voltage Eo is light (Po=25 W). In particular, the switching frequency decreases in response to a decrease in the level of the secondary side DC output voltage Eo as the load becomes heavier, but the switching frequency increases in response to an increase in the level of the secondary side DC output voltage Eo as the load becomes lighter. This indicates the fact that a constant voltage controlling operation by upper side control is performed as a switching frequency controlling method.
Further, since the operation on the primary side described above is obtained, an alternating voltage V2 having the waveform shown in FIG. 46 is induced in the secondary winding N2 of the insulating converter transformer PIT. Then, within the period of any one of the half cycles in which the alternating voltage V2 has a positive polarity, the rectification diodes [Do1, Do4] on the secondary side conduct to allow rectification current ID1 to flow in the waveform and at the timing shown in FIG. 46. Further, within the period of the other one of the half cycles in which the alternating voltage V2 has a negative polarity, the rectification diodes [Do2, Do3] on the secondary side conduct to allow rectification current ID3 to flow in the waveform and at the timing shown in FIG. 46. Further, as seen in FIG. 46, the rectification currents ID1 and ID3 are composed to form secondary winding current I2 to flow to the secondary winding N2.
FIG. 47 illustrates an AC→DC power conversion efficiency and a characteristic of the switching frequency of the power supply circuit shown in FIG. 44 with respect to the load variation under the input voltage condition of the AC input voltage VAC=100 V.
The switching frequency fs has a characteristic that it decreases as the load increases because the constant voltage controlling operation is performed. However, the characteristic just described is not a linear variation characteristic with respect to the load variation. For example, within a range from the load power Po=approximately 25 W to Po=0 W or less, the tendency is exhibited that the switching frequency fs increases steeply.
Meanwhile, the AC→DC power conversion efficiency (ηAC→DC) has a tendency that it increases as the load power Po increases, and when the load power Po=150 W, an ηAC→DC power conversion efficiency of approximately 91.0% is obtained.
It is to be noted that, in order to obtain the experimental results described with reference to FIGS. 46 and 47, the components shown in FIG. 44 are set as mentioned just below.    Insulating converter transformer PIT(EER-35 type ferrite core, gap length=1.4 mm, coupling coefficient k=0.75)Primary winding N1=35 T (turns), secondary winding N2=50 T    Primary side series resonance capacitor C1=0.039 μF    Partial resonance capacitor Cp=330 pF
Another example of the switching power supply circuit is shown in the circuit diagram of FIG. 45. It is to be noted that, in FIG. 45, like elements to those of FIG. 44 are denoted by like reference characters, and a description of them is omitted herein to avoid redundancy.
The secondary side rectification circuit of the power supply circuit shown in FIG. 45 includes a full wave rectification circuit. In particular, a center tap is provided for the secondary winding N2 such that the secondary winding N2 is divided into secondary winding sections N2A and N2B. In this instance, the secondary winding sections N2A and N2B are formed from numbers of turns equal to each other. Further, the center tap is grounded to the secondary side ground. Furthermore, the rectification diodes Do1 and Do2 and the secondary side smoothing capacitor Co are connected to the secondary winding N2. By the full wave rectification circuit, the secondary side DC output voltage Eo can be obtained as a voltage across the smoothing capacitor Co.
A power supply circuit having the configuration described above may be provided as the power supply of a plasma display apparatus. In the plasma display apparatus, the load power Po varies over a comparatively wide range, for example, from Po=100 W or more to no load. Further, a secondary side DC output voltage having, for examples a comparatively high level of 200 V or more is required.
When an experiment regarding the power supply circuit shown in FIG. 45 was performed, results of operation and a characteristic substantially equal to those illustrated in FIGS. 46 and 47 were obtained.
It is to be noted that, when the experiment was performed, the components of the circuit shown in FIG. 45 were set as given below.    Insulating converter transformer PIT(EER-35 type ferrite core, gap length=1.4 mm, coupling coefficient k=0.75)Primary winding N1=35 T (turns), secondary winding N2=secondary winding section N2A+secondary winding N2B=50 T+50 T=100 T    Primary side series resonance capacitor C1=0.039 μF    Partial resonance capacitor Cp=330 pF
As described above, the power supply circuit shown in FIG. 45 is provided as a power supply of a plasma display apparatus and is configured so that the secondary side DC output voltage Eo obtained has a comparatively high level. In order to cope with this, in the circuit shown in FIG. 45, the secondary side rectification circuit is formed as a full wave rectification circuit, and the number of turns of the secondary winding N2 is suitably increased to 100 T.
Incidentally, where the configuration as a resonance type converter for implementing stabilization of the secondary side DC output voltage by the switching frequency controlling method is adopted as in the case of the power supply circuit shown in FIG. 44 (FIG. 45), the adjustable controlling range of the switching frequency for stabilization is a comparatively wide range.
This is described with reference to FIG. 48. FIG. 48 illustrates a constant voltage controlling characteristic of the power supply circuit shown in FIG. 44 (FIG. 45) in the form of a relationship between the level of the switching frequency fs and the level of the secondary side DC output voltage Eo.
It is to be noted that, in the description given with reference to FIG. 48, it is a premise that the power supply circuit in FIG. 44 (FIG. 45) adopts upper side control as the switching frequency controlling method. Upper side control as used herein is a controlling method for adjustably controlling the switching frequency within a frequency range higher than a resonance frequency fo1 of the primary side series resonance circuit and utilizing the variation of the resonance impedance caused by the adjustment control to control the level of the secondary side DC output voltage Eo.
Generally, the resonance impedance of the series resonance circuit is lowest at the resonance frequency fo1. Consequently, as a relationship between the secondary side DC output voltage Eo and the switching frequency fs in upper side control, the level of the secondary DC output voltage Eo increases as the switching frequency fs approaches the resonance frequency fo1, but decreases as the switching frequency fs moves away from the resonance frequency fo1.
Accordingly, as seen in FIG. 48, the level of the secondary side DC output voltage Eo with respect to the switching frequency fs in a condition that the load power Po is constant exhibits a quadratic curve variation. In particular, the level of the secondary side DC output voltage Eo exhibits a peak when the switching frequency fs is equal to the resonance frequency fo1 of the primary side series resonance circuit, and decreases as the switching frequency fs moves away from the resonance frequency fo1.
Further, the level of the secondary side DC output voltage Eo corresponding to the switching frequency fs in the same condition as that described above exhibits a characteristic that it shifts such that the level at the maximum load power Pomax is less by a predetermined amount than the level at the minimum load power Pomin. In particular, where it is considered that the switching frequency fs is fixed, the level of the secondary side DC output voltage Eo decreases as the load condition becomes heavier.
If an attempt is made to stabilize the secondary side DC output voltage Eo by upper side control so that Eo=tg may be satisfied where such a characteristic as just described is exhibited, then the adjustment range (necessary control range) of the switching frequency necessary for the power supply circuit shown in FIG. 44 (FIG. 45) is a range indicated by the reference character Δ fs.
Actually, the power supply circuit shown in FIG. 44 performs the constant voltage control based on the switching frequency controlling method so that the secondary side DC output voltage Eo is stabilized at 135 V so as to cope with the input variation range and the load conditions. The variation range is from the AC input voltage VAC=85 V to 120 V of the AC 100 V type. The load conditions are the maximum load power Pomax=150 W and the minimum load power Pomin=0 W (no load) to the secondary side DC output voltage Eo, which is a main DC power supply.
In this instance, the variation range of the switching frequency fs that is varied for the constant voltage control by the power supply circuit shown in FIG. 44 is fs=80 kHz to 200 kHz or more, and also the range Δ fs is a correspondingly wide range of 120 kHz or more.
Further, the power supply circuit shown in FIG. 45 performs the constant voltage control so that the secondary side DC output voltage Eo is stabilized at the rated level of approximately 200 V. Therefore, similar to the power supply circuit shown in FIG. 44, the range Δ fs of the power supply circuit shown in FIG. 45 is a correspondingly wide range.
As one of the power supply circuits, a power supply circuit ready for a wide range is known, which is configured so as to operate with an AC input voltage range, for example, from approximately AC 85 V to 288 V. It can be applied both in an area in which the AC input voltage of the AC 100 V type is used, such as, for example, Japan, U.S.A. and so forth, and in another area in which the AC input voltage of the AC 200 V type is used, such as, for example, Europe.
Thus, it is considered here that the power supply circuit shown in FIG. 44 (FIG. 45) is configured as a power supply circuit ready for a wide range as described above.
As described above, where the power supply circuit is ready for a wide range, it is ready for an AC input voltage range, for example, from AC 85 V to 288 V. Accordingly, the variation range of the level of the secondary side DC output voltage Eo increases when compared with that in an alternative case in which the power supply circuit is ready for a single range of, for example, only the AC 100 V type or only the AC 200 V type. In order to perform the constant voltage control for the secondary side DC output voltage Eo whose variation range is expanded in correspondence to such an AC input voltage range as described above, switching frequency control over a still wider range is required. For example, in the power supply circuit shown in FIG. 44 (FIG. 45), it is necessary to expand the controlling range of the switching frequency fs to approximately 80 kHz to 500 kHz.
However, the upper limit to the driving frequency with which an IC (oscillation driving circuit 2) for driving an actual switching device can cope is approximately 200 kHz. Further, even if a switching driving IC that can be driven with such a high frequency as described above is configured and mounted, when the switching device is driven with such a high frequency as described above, the power conversion efficiency drops remarkably. Therefore, it is difficult to practically use the switching driving IC described above as an actual power supply circuit. Incidentally, the upper limit to the AC input voltage VAC which can be stabilized, for example, by the power supply circuit shown in FIG. 44 (FIG. 45), is approximately 100 V.
Therefore, in order to configure a switching power supply circuit using switching frequency control for stabilization as a switching power supply circuit ready for a wide range, it is known to adopt, for example, such a configuration as described below.
In particular, a rectification circuit system receiving a commercial AC power supply to produce the DC input voltage (Ei) is provided with a function of performing a changeover between a voltage doubler rectification circuit and a full wave rectification circuit in response to an input of commercial AC power supply of the AC 100 V type and the AC 200 V type.
In this instance, the circuit is configured such that the level of the commercial AC power supply is detected and the circuit connection of the rectification circuit system is changed over by a switch using an electromagnetic relay such that a voltage doubler rectification circuit or a full wave rectification circuit is formed in response to the detected level.
However, in such a configuration for the changeover of the rectification circuit system as described above, a number of electromagnetic relays are required as described above. Further, at least a pair of smoothing capacitors must be provided in order to form a voltage doubler rectification circuit. Therefore, the number of parts increases and this increases the cost. Also, the mounting area of a power supply circuit board is expanded to increase the size of the circuit. In particular, since the smoothing capacitors and the electromagnetic relays are large-size parts from among those parts forming the power supply circuit, the size of the board becomes rather large.
Where the configuration for changeover between full wave rectification operation and voltage doubler rectification operation is applied, if, when the commercial AC power supply of the AC 200 V type is input, the level of the AC input voltage becomes lower than that corresponding to the AC 200 type because an instantaneous service interruption occurs or the AC input voltage decreases to a voltage level lower than the rated voltage or the like, then a malfunction may occur in which it is detected that the commercial AC power supply input is the AC 100 V type and a changeover from the full wave rectification circuit to the voltage doubler rectification circuit is performed. If such a malfunction occurs, then the voltage doubler rectification will be performed for an AC input voltage having the level of the AC 200 V type. Therefore, there is the possibility that, for example, the switching devices Q1, Q2 and so forth may be broken by being subjected to a voltage higher than that which they can withstand.
Therefore, as an actual circuit, in order to prevent the occurrence of such a malfunction as described above, a configuration is adopted in which not only is the DC input voltage of the switching converter on the main power supply side detected, but the DC input voltage of the converter circuit on the standby power supply side is also detected. Consequently, a member for detecting the converter circuit on the standby power supply side must be added, and an increase of the cost described above and an increase of the size of the circuit board result.
Further, the DC input voltage of the converter on the standby power supply side is detected in order to prevent a malfunction. This signifies that only an electronic apparatus which includes not only a main power supply but also a standby power supply can actually use a power supply circuit which includes a circuit for changing over the rectification operation and which is ready for a wide range. In other words, the type of electronic apparatus in which the power supply can be incorporated is limited to that which includes a standby power supply, and the range of utilization becomes much narrower.
Further, as a configuration ready for a wide range, a configuration is also known in which the form of the current resonance converter on the primary side is changed over between that of a half bridge connection and that of a full bridge connection in response to an input of a commercial power supply of the A 100 V type/AC 200 V type.
With the configuration just described, even if the AC input voltage of the AC 200 V type drops to the level of the AC 100 V type, for example, as a result of such instantaneous interruption as described above or the like to cause a malfunction, the switching operation is only changed over from a half bridge operation to a full bridge operation. As a result, a situation does not arise in which a voltage higher than the withstanding voltage is applied to the switching devices. Therefore, the DC input voltage on the standby power supply side need not be detected. Consequently, the configuration can be applied to an electronic apparatus that does not include a standby power supply. Further, since a changeover of the commercial power supply line is not involved, a changeover of the circuit formation by a semiconductor switch is possible. Therefore, a large-size switching member such as an electromagnetic relay need not be provided.
However, with the configuration described above, at least four switching devices must be provided in order to form a full bridge connection in response to the AC 100 V type. In particular, in comparison with the configuration of a converter using only a half bridge connection, which can be formed from two switching devices, an additional two switching devices must be added.
Further, in the configuration described, four switching devices perform the switching operation in the full bridge operation, and three switching devices perform the switching operation even in the half bridge operation. While the resonance converter produces low switching noise, the disadvantage in regard to switching noise increases as the number of switching devices performing switching operations in such a manner as described above increases.
Also, where any one of the configurations described above is adopted as a configuration ready for a wide range in such a manner as described above, when compared with an alternative configuration ready for a single range, an increase in the circuit scale and an increase in the cost caused by an increase in the number of parts cannot be avoided. Further, intrinsic problems which do not appear with the configuration ready for a single range, such as a limitation on the range of apparatuses which can be utilized and an increase in the switching noise and so forth occur with the former configuration and the latter configuration, respectively.
Further, where the control range of the switching frequency is a suitably wide range as in the case of the power supply circuit shown in FIG. 44 (FIG. 45), a problem also occurs that a high-speed response characteristic in the stabilization of the secondary side DC output voltage Eo decreases.
Some electronic apparatus involve varying operations such that the load condition changes over instantaneously, for example, between a state in which the load has a maximum level and another state in which the load is substantially zero. A load exhibiting such a load variation as just described is also called a switching load. The power supply circuit to be incorporated in such an apparatus as just described must be configured so that the secondary side DC output voltage is appropriately stabilized against a load variation such as that of a switching load.
However, as described above with reference to FIG. 48, where the switching frequency has a characteristic of a wide control range, a comparatively long time is required to adjust the switching frequency with which the secondary side DC output voltage is provided to the required level in response to a load variation such as that of the switching load described above. In short, an undesirable result is obtained as the response characteristic of the constant voltage control.
It can be recognized that the power supply circuit shown in FIG. 44 (FIG. 45) is particularly disadvantageous in the constant voltage control response characteristic to a switching load as described above. In the switching frequency characteristic according to the constant voltage control, the switching frequency varies by a great amount within the load range from the load power Po=approximately 25 W to 0 W as seen in FIG. 47.